Signal processing circuit for spread spectrum communications

ABSTRACT

In a transmitting unit, an input signal is converted into an I data sequence of an in-phase (I) symbol and a Q data sequence of a quadrature (Q) symbol, a differential encoding is performed for the I and Q data sequences according to a differential binary phase shift keying (DBPSK) modulation to produce an I code sequence and a Q code sequence, a spectrum of the I code sequence is spread by a first spreading code signal to produce an I spread spectrum signal, a spectrum of the Q code sequence is spread by a second spreading code signal to produce a Q spread spectrum signal, the I and Q spread spectrum signals are quadrature-modulated to a synthesized signal, and the synthesized signal is transmitted to a receiving unit. In the receiving unit, the synthesized signal is quadrature-demodulated to a first baseband signal and a second baseband signal, the first and second baseband signals are respectively correlated with each of the first and second spreading code signals to obtain first and second I correlated signals and first and second Q correlated signals, and a BPSK delay detection is performed for a group of the first and second I correlated signals and a group of the first and second Q correlated signals to reproduce the I and Q data sequences.

This application is a division of application Ser. No. 08/551,111 filedOct. 31, 1995, now U.S. Pat. No. 5,610,939, issued Mar. 11, 1997, whichis a division of application Ser. No. 08/197,592 filed Feb. 17, 1994 nowU.S. Pat. No. 5,488,629, issued Jan. 30, 1996.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to signal processing circuits for performing asynchronization acquisition and a frequency (offset) correction in caseof using a spread quadrature modulation wave in a spread spectrumcommunication system. Further, this invention relates to a digitalcorrelator for use in a receiving component of a spread spectrumcommunication system of the direct sequence modulation type. Moreover,this invention relates to an M phase shift keying (hereunder abbreviatedas M-PSK (incidentally, M>4)) modulation/demodulation method employed ina spread spectrum communication system of the direct sequence modulationtype. Furthermore, this invention relates to a spreading-code generatingmethod for use in a code division multiple access (hereunder abbreviatedas CDMA) system employing a spread spectrum communication method of thedirect sequence modulation type.

2. Description of the Related Art

In recent years, it has been studied how a communication network usingradio techniques (for example, a cellular network or a local areanetwork (LAN)) is put to practical use. As an example of a communicationmethod employed in such a communication network, a CDMA method using aspectrum spreading has been studied.

Further, two principal types of the spectrum spreading are a directsequence (hereunder abbreviated as DS) type and a frequency hopping(hereunder abbreviated as FH) type. The DS type of spectrum spreading(hereunder sometimes referred to as DS/CDMA) is a method of effectingcommunications by directly performing a spectral spreading on aninformation signal by use of a spreading code or pattern having afrequency which is far higher than the frequency of the informationsignal (for instance, a frequency which is tens to thousands times thefrequency of the information signal). The FH type of spectrum spreading(hereunder sometimes referred to as FH/CDMA) is a method of spreadingthe spectrum of a narrowband frequency-modulated signal by changing thefrequency of a carrier according to a spreading-code pattern andaveraging the signal resultingly. Incidentally, a CDMA method is acommunication method for performing a multiplexing within a samefrequency band by using different spreading-code patterns when effectinga spreading of the DS or FH type.

Previously, a binary phase shifting keys (hereunder abbreviated as BPSK)has been mainly used in case of the DS type of spectrum spreading.Recently, for the purpose of performing data communication at a highspeed, a quadrature phase shift keying (hereunder abbreviated as QPSK)by which spectrum spreading and synthesization are performed on each ofan in-phase channel (hereunder abbreviated as Ich) and a quadrature(namely, 90°-out-of-phase) channel (hereunder abbreviated as Qch), aswell as an M-PSK (incidentally, M>4), has become studiedenthusiastically.

In case of employing the spectrum spreading, the influence of what iscalled a transmission/reception frequency offset of a carrier wave atthe time of initial synchronization acquisition is a serious matter. Thereason is as follows. Namely, in cases of other communication systems(for instance, an FM broadcasting system and an analog automobiletelephone system), a transmission/reception frequency offset can besuppressed by effecting a tracking by use of an automatic frequencycontrol (hereunder abbreviated as AFC) in such a manner to maximize thesignal level of a reception signal (namely, a received signal). Incontrast, in case of employing the spectrum spreading, the signal levelof a desired reception signal is sometimes lower than a noise leveluntil the initial acquisition is completed and thus an AFC as used inthe other communication systems cannot be realized in case of acommunication system employing the spectrum spreading. Therefore, amethod of sweeping the frequency of a carrier within a maximum frequencyoffset range basically is used in an AFC of the system employing thespectrum spreading. Such a method, however, has a drawback in that ittakes time to perform a synchronization acquisition.

Thus a method of calculating the envelopes of components of a receptionsignal regardless of a frequency offset is employed in the conventionalsystem using BPSK.

FIG. 1 is a schematic block diagram for illustrating the configurationof a conventional synchronization acquisition circuit of BPSK type. Inthis figure, reference numeral 310 designates a radio antenna; 320 aband-pass filter (hereunder abbreviated as BPF); 330 an AGC circuit; 341and 342 down-mixers corresponding to Ich and Qch, respectively; 351 alocal signal source; 352 a (π/2)-phase shifter; 361 and 362 low-passfilters (hereunder abbreviated as LPFs); 371 and 372 analog-to-digital(hereunder referred to simply as A/D) converters: 381 and 382correlating detectors (hereunder sometimes referred to as correlators)for what is called a de-spreading; 391 and 392 squaring devices; 400 anadder; 410 a synchronization acquisition judgement circuit; 420 a datadecoding circuit; 430 a spreading-code generating circuit; and 500 acode clock recovery circuit.

In the synchronization acquisition circuit of FIG. 1, a quadraturedetection of a reception signal is first performed to obtain thechannels (or components) Ich and Qch. Then, a correlating detection isperformed on each of the components by using the same spreading code(namely, the sequence of the same spreading code words). Subsequently,the square of an output of each of the correlating detectors isobtained. In addition, the obtained squares of the outputs of thecorrelating detectors are added up to obtain the magnitude of theenvelope (to be described later) represented on a phase plane. Thereby,the influence of the frequency offset at the time of synchronizationacquisition is eliminated.

The above described operation of this synchronization acquisitioncircuit will be further explained hereinbelow by using expressions orequations (1) to (12). Here, let Dn, C, ω.sub.∘, Δω and N denote theamplitude of an information signal, a spreading code, a carrier angularfrequency, a transmission/reception frequency offset and a receptionin-band noise power. Further, the spreading code is C={c.sub.□, c_(i) .. . c_(M-1) } (incidentally, M is a period). Here, a reception spectrumspreading signal y(i) is assumed to be given by the following equation:

    y(i)=D.sub.n c.sub.i cos (ω.sub.∘ +Δω)t+D.sub.n c.sub.i sin (ω.sub.∘ +Δω)t+N                                       (1)

where N is assumed to be given by the following equation:

    N=N.sub.i cos ω.sub.∘ t+N.sub.q sin ω.sub.∘ t                               (2)

First, a quadrature detection is performed on the signal represented bythe equation (1) (namely, the equation (1) is divided by exp(jω.sub.∘t)) to obtain the in-phase channel Ich and the quadrature channel Qch.Then, these channels (or components) are applied to the LPFs,respectively. Thus output signals r(Ich) and r(Qch) of the LPFscorresponding to the input components Ich and Qch are obtained asfollows:

    r(Ich)=D.sub.n c.sub.i cos Δωt+N.sub.i         ( 3)

    r(Qch)=D.sub.n c.sub.i sin Δωt+N.sub.q         ( 4)

Subsequently, a correlating detection is performed on each of thesignals represented by the equations (3) and (4) by using a spreadingcode C' which is similar to the spreading code C but may be differentfrom the sequence C only in phase. As is apparent from the definition ofthe spreading code, an output of each of the correlating detectors canbe equal to or greater than a predetermined value only in case whereC'=C. If the phase due to the frequency offset can be regarded asconstant for one period of the spreading code C', the signal levelscorresponding to the channels Ich and Qch are obtained as represented bythe following equations (5) and (6), respectively:

    Σ{r(Ich)×c'.sub.i } MD.sub.n cos Δωt(5)

    Σ{r(Ich)×c'.sub.i } MD.sub.n sin Δωt(6)

Then, the squares of the right sides of the equations (5) and (6) areadded up as follows: ##EQU1##

As is seen from the equation (7), the frequency offset Δω vanishes andthe square of the magnitude MD_(n) of the envelope is obtained. Thus thejudgement on the synchronization acquisition can be effected.

However, in case where QPSK is employed in the conventional system, ifthe spreading codes C_(I) and C_(Q) corresponding to Ich and Qch,respectively, are different from each other and Δω becomes equal to(π/2) due to the influence of the frequency offset, a correlation cannotbe detected. Namely, the reception signal y(i) in case of this case isassumed to be represented by the following equation:

    y(i)=D.sub.In c.sub.Ii cos (ω.sub.∘ +Δω)t+D.sub.In c.sub.Qi sin (ω.sub.∘ +Δω)t+N                                       (8)

Then, a quadrature detection is performed on this reception signal(namely, the equation (8) is divided by exp(jω.sub.∘ t)) to obtain thein-phase channel Ich and the quadrature channel Qch. Subsequently, thesechannels (or components) are inputted to the LPFS, respectively. Thusoutput signals r(Ich) and r(Qch) of the LPFs corresponding to the inputchannels or components Ich and Qch are obtained as follows:

    r(Ich)=D.sub.In c.sub.Ii cos Δωt-D.sub.Qn c.sub.Qi sin Δωt+N.sub.i                                   ( 9)

    r(Qch)=D.sub.Qn c.sub.Qi cos Δωt+D.sub.In c.sub.Ii sin Δωt+N.sub.q                                   ( 10)

Here, the equations (9) and (10) can be rewritten as follows bysubstituting (π/2) for Δω.

    r(Ich)=-D.sub.Qn c.sub.Qi sin Δωt+N.sub.i      ( 11)

    r(Qch)=D.sub.In c.sub.Ii +N.sub.q                          ( 12)

Thus, before the correlation detection, the component including c_(Ii)is removed from the signal r(Ich) corresponding to Ich. Further, thecomponent including c_(Qi) is removed from the signal r(Qch)corresponding to Qch.

Therefore, in this case, the influence of a frequency offset can not beeliminated if the process effected in case of employing BPSK isperformed. Consequently, it is difficult to achieve a synchronizationacquisition. The present invention is created to resolve such a drawbackof the conventional system.

It is, therefore, an object of the present invention to provide a signalprocessing circuit which can make a synchronization acquisition judgmentwithout being influenced by a frequency offset.

Meanwhile, as the result of the recent progress in LSI techniques or thelike, a spread spectrum communication system has come to be applied notonly to military or satellite communications but also to industrial orprivate equipment. Especially, the application of the spread spectrumcommunication system to a cellular type mobile communications is nowstudied in the United States and so forth. Thus the spread spectrumcommunication techniques have rapidly come to draw the attention of theworld. Among the various types of the spread spectrum communicationmethods, the direct sequence modulation method for spreading data by useof spreading-codes referred to as spread signals is advantageous in thata system for performing such a method can be constructed easily by usingLSIs and that a distance can be measured by checking a time required todetect a correlation. Thus many research institutes proceed with thedevelopment of the direct sequence modulation method.

Hereinafter, a conventional spread spectrum communication system of thedirect sequence modulation type will be described briefly.

FIG. 2(a) is a schematic block diagram for illustrating theconfiguration of a spreading circuit of the conventional spread spectrumcommunication system. Further, FIG. 2(b) is a schematic block diagramfor illustrating the configuration of a de-spreading circuit of theconventional spread spectrum system. In these figures, referencenumerals 901, 902 and 909 are 2-input multipliers; 903 and 914spreading-code generators for generating spreading-codes; 904 and 910local oscillators for outputting local oscillation signals to beconverted by the multipliers 902 and 909 into signals having radiofrequencies or intermediate frequencies; 905 a power amplifier foramplifying signals of radio frequencies and transmitting the amplifiedsignals; 906 and 907 a transmitting antenna and a receiving antenna; 908a receiving front-end circuit for selecting components havingfrequencies of a required band from received signals of radiofrequencies and increasing the levels of the selected components to anecessary signal level; 911 an A/D converter for converting a signalhaving a frequency of a baseband, which is obtained by the multiplier909, to a digital signal; 912 a correlator; 913 a synchronizing circuitfor monitoring a synchronization acquisition state according to acorrelation output signal of the correlator; and 915 a clock generatorfor generating timing clock signals for the A/D converter 911, thespreading-code generator 914 and so on according to information obtainedby the synchronizing circuit 913.

The circuit of FIG. 2(a) is a spreading portion of a transmitting unit.Further, a data signal to be transmitted is inputted to the multiplier901 from left, as viewed in this figure. Data represented by the datasignal is multiplied by a code represented by a spreading-code signal,which is generated in the spreading-code generator 903, by themultiplier 901. Namely, the multiplier 901 outputs a signal, thespectrum of which is spread over the frequencies of the spreading-codesignal. Here, pseudo-noise code (PN code) signals are used as thespreading code signals (incidentally, typical examples of a PN code(set) are what is called the M-code and what is called the Gold code).(Incidentally, the autocorrelation characteristics of each ofspreading-code patterns and the characteristics of the cross correlationbetween a spreading-code pattern of a code sequence and anotherspreading-code pattern of the same code sequence vary with the code.)The spectrum of a data signal to be transmitted is spread by using thisspreading-code signal. Further, the spreading of a data signal issometimes effected at a frequency 2^(n) -times the frequency of the datasignal, for the ease of de-spreading and for the simplicity of theconfiguration of the circuit.

Next, the signal spread by the multiplier 901 is mixed by the multiplier902 with a signal sent from the local oscillator 904. Then, a resultantsignal is amplified by the power amplifier 905. Thereafter, theamplified signal is transmitted from the antenna 906.

In contrast, in a receiving unit of FIG. 2(b), a de-spreading iseffected by performing a reverse procedure of the spreading (namely,performing a demodulation) and the original signal (namely, the datasignal) is recovered. First, a signal obtained by increasing the signallevels of a part of a signal received from the antenna 907, which partcorresponds to a required band, to a necessary signal level ismultiplied by a local oscillation signal of the same frequency as theoscillation frequency of the local oscillator of the transmitting unit,which is issued from the local oscillator 910, in the multiplier 909.Thus a baseband signal, which is spread by using the spreading-code, isobtained. Subsequently, such a signal is converted by the A/D converter911 into a digital signal. Then, the correlator 912 obtains acorrelation value from this digital signal and another signal sent fromthe spreading-code signal generator 914 which generates the samespreading-code signal as generated in the transmitting unit.

Next, the synchronizing circuit 913 monitors the synchronizationacquisition state according to an output of the correlator 912 andcontrols the spreading-code signal generator 914 and the clock generator915 to provide a feedback to the correlator 912. Namely, a feedback loopis established in this process. Thus, the circuit 913 operates to stablyobtain an output of the correlator and ensure the synchronization.

FIG. 3 is a detail block diagram for illustrating the configuration ofthe conventional correlator which employs a digital matched filterpractically. In this figure, reference numeral 101 designates a shiftregister for shifting data represented by a signal, which is obtained asa result of the A/D conversion, in response to each sampling clock. Thisshift register has capacity sufficient to store data sent thereto forwhat is called a spreading period. Further, reference numeral 102denotes an arithmetic circuit for producing a product of aspreading-code and data represented by a signal inputted to the shiftregister 101; and 103 an adder for obtaining a total sum of results ofarithmetic operations effected in the arithmetic circuit 102.

In this conventional correlator with the configuration describedhereinabove, a signal obtained by performing A/D conversion on thereception signal converted into a baseband signal is inputted to theshift register 101. As shown in this figure, a product of each ofsignals γ.sub.□ to γ.sub.(n-1) inputted to the shift register 101 and acorresponding one of spreading-code signals ref.sub.□ to ref.sub.(n-1)is calculated by the arithmetic circuit 102. Further, a correlationoutput or value is obtained from a result of the addition effected bythe adder 103. In case where 1 bit of the transmitted data issynchronized with the spread signal correspondingly to one period of thespreading-code signal, the data can be decoded by making comparisonsamong the correlation values, for example, by using the maximum andminimum values of the correlation outputs. Further, the frequency of aclock signal can be regulated such that the square of the correlationvalue maintains its maximum value. Moreover, the same number as of thecorresponding spreading-codes or an integral multiple thereof for aclock correction is often selected as the number of stages of shiftregisters.

The conventional correlator, however, has drawbacks in that a largenumber of gates such as a shift register, a multiplier and an adder areneeded and thus the size of the circuit becomes large and that the powerconsumption thereof also becomes large. Especially, in case of theconventional correlator of the type that performs a sampling of 1 bit ofthe spreading-code signal a plurality of times, the operating frequencyof the adder 103 becomes high. This drawback affects the realizabilityof the conventional correlator of such a type. The present invention isaccomplished to eliminate the drawbacks of the conventional correlator.

It is, accordingly, an object of the present invention to provide adigital correlator which can reduce the size and power consumptionthereof and increase the operating frequency thereof.

FIG. 4(a) is a schematic block diagram for illustrating the transmittingunit of a conventional spread spectrum communication system of thedirect sequence type that employs a QPSK. Further, FIG. 4(b) is aschematic block diagram for illustrating the receiving unit of thisspread spectrum communication system.

In the transmitting unit of FIG. 4(a), an input signal is first inputtedto a QPSK encoder 1101 which performs a mapping of the input signal ontosymbol signals representing symbols used in QPSK modulation. Thus datarepresented by the input signal are converted into two data sequencescorresponding to symbols I and Q, respectively. These data sequences Iand Q are multiplied by data represented by signals sent fromcorresponding spreading-code generators 1104_(i) and 1104_(q) inmodulo-2 multipliers 1103_(i) and 1103_(q), respectively. Then, resultsof the multiplications are sent therefrom to a quadrature modulationportion 1109 in which PN-code signal is often used as a spreading-codesignal. The spectrum of a signal representing data to be transmitted isspread by using the spreading-code signal. In this figure, thequadrature portion 1109 is indicated by being surrounded by dashedlines. Further, in mixers 1105_(i) and 1105_(q) of the portion 1109, therespective output signals of the generators 1104_(i) and 1104_(q) aremixed with a signal sent from a first local oscillator 1107 and anothersignal obtained by shifting the phase thereof by (π/2). Then, outputs ofthe mixers 1105_(i) and 1105_(q) are added up by an adder 1108.Thereafter, an output signal of the adder 1108 is mixed by a mixer 1110with a signal sent from a second local oscillator 1111 and thus isconverted into a signal of a carrier band. Finally, an output signal ofthe mixer 1110 is transmitted from an antenna 1112.

In the receiving unit of FIG. 4(b), a signal received from an antenna1113 is first mixed by a mixer 1114 with another signal sent from afirst local oscillator 1115 and is thus converted into a signal of anintermediate frequency band. Then, the converted signal is furtherconverted by mixers 1116_(i) and 1116_(q) into baseband signalscorresponding to I and Q, respectively, by using a signal sent from asecond local oscillator 1119 and another signal obtained by a phaseshifter 1117 by shifting the phase thereof by (π/2). Such a frequencyconversion of a signal to baseband signals by using quadrature signalsis called as a quadrature detection. Further, a quadrature detectionportion 1118 consists of the mixers 116_(i) and 1116_(q) and the phasesifter 1117. Thereafter, correlation values are obtained from datasignals (namely, the baseband signals) and spreading-code signalsgenerated by spreading-code generators 1121_(i) and 1121_(q), which arethe same as those generated by the spreading-code generators of thetransmitting unit. It is, however, rare that the sum of the oscillationfrequencies of the first and second local oscillators 1115 and 1119 ofthe receiving unit is exactly equal to the carrier frequency of thetransmitting unit (namely, the sum of the oscillation frequencies of thefirst and second local oscillators of the transmitting unit). Thus whatis called a phenomenon of "(phase) rotation" of data on a phase plane(namely, a phase shift of data) is liable to take place. Therefore, inorder to decode data, a correlation is obtained by using 2 correlators1120a to 1120d. Further, an angle of "phase rotation" (namely, an amountof angular displacement or phase shift) is obtained by a phase detectingcircuit 1701. Finally, data is decoded, performing a frequency offsetcompensation in a QPSK data decoder 1702.

This conventional system, however, has drawbacks in that theconfigurations of the phase detecting circuit and the QPSK data decoderbecome complex and that if the transmission rate becomes high, thecircuit thereof is limited in processing speed. The present invention iscreated to eliminate these drawbacks of the conventional system.

It is, therefore, an object of the present invention to provide an M-PSKmodulation/demodulation method, by which the configuration of a decodingportion can be simplified without spreading a necessary band in casewhere a transmission rate is the same as of the conventional system.

It is another object of the present invention to provide an M-PSKmodulation/demodulation method, by which good characteristics can beobtained without a frequency-offset compensation circuit.

It is a further object of the present invention to provide an M-PSKmodulation/demodulation method, by which characteristics of a spreadspectrum communication system can be improved when effecting amultiplexing.

It is still another object of the present invention to provide an M-PSKmodulation method, by which the processing speed of a signal processingcircuit of a spread spectrum communication system can be increased incomparison with the conventional system employing a QPSK modulation.

Meanwhile, in a conventional system employing a CDMA method, amultiplexing is performed on signals of a same frequency band in orderto increase channel capacity. This conventional system, however, hasencountered a problem in that such a multiplexing is difficult whenusing only the M-code or the Gold code.

Further, it is preferable for suppressing the interference betweenspread waves or signals to use spreading-code patterns having smallcross correlation. However, the kinds of the combination of suchspreading-code patterns are limited. Moreover, in case of a system ofthe DS type employing a QPSK (hereunder sometimes referred to as DS/QPSKtype), different spreading-code patterns are used corresponding to Ichand Qch, respectively. However, in case where two channels (orcomponents) Ich and Qch are not completely separated in a receiving unitdue to a frequency offset after a quadrature detection, if the crosscorrelation between the spreading-codes respectively corresponding tothe channels (or components) Ich and Qch is large, the interferencebetween the components occurs at the time of performing a correlationdetection on each of the components Ich and Qch.

As to the problem relating to the channel capacity, what is called thedegree of multiplexing (hereunder sometimes referred to as themultiplexing degree) can be increased by lengthening the period of thespreading-code to increase what is called a spreading rate. However, ina practical system, the spreading rate is limited due to the conditionssuch as the relation between the spreading band width and theinformation transmission rate and the operating speed of the system.Thus, practically, it is difficult to employ the spreading code of asufficiently long period.

In case of employing a CDMA method, the relation between the degree ofmultiplexing and the quality of communication depends on thecharacteristics of the cross correlation between the code patterns to alarge extent. As above stated, the M-code, the Gold code sequence or thelike have been studied as the spreading code for a spectrum spreading.For example, it is known that the M-code has good autocorrelationcharacteristics and thus the peak of the correlation value can be easilydetected, though the number of code patterns which can be generatedtherefrom is small. In contrast, in case of the Gold code, the number ofcharacter patterns which can be generated therefrom is larger than thatin case of the M-code. However, the Gold code has undesirablecharacteristics of the cross correlation between code patterns thereof.Thus, when performing a multiplexing, the quality of communication isextremely deteriorated due to the interference between the spread wavesor signals. Hence, there is limitation on the number of channels whichcan be used for communication simultaneously.

Recently, there has been studied a method using an orthogonal code whichhas good characteristics of the cross correlation between code patternsobtained therefrom. Generally, in case where the orthogonality betweencode patterns of the orthogonal code is maintained, there is no crosscorrelation therebetween and thus the degree of multiplexing can belarge. In contrast, when the orthogonality is lost, the crosscorrelation between the code patterns thereof is deterioratedconsiderably. Therefore, what is called an intersymbol synchronizationis necessary in case of employing the orthogonal code in a CDMA systemas the spreading code.

Further, in case of employing a Hadamard sequence which is a kind oforthogonal code, a Hadamard matrix having the code length of 2.sup.(n+1)is generated by performing a Hadamard transform on a 2×2 Hadamard matrixn times. As can be seen from this generation process, a code patternconsists of repetitive code sub-patterns. Thus, a Hadamard code hasundesirable characteristics of the autocorrelation. Consequently, aconventional system employing a Hadamard code has a drawback in that itis difficult to achieve a synchronization acquisition and a multipathseparation. The present invention is accomplished to eliminate thisdrawback of the conventional system.

It is, accordingly, an object of the present invention to provide anM-PSK modulation method by which the interference between the phasecomponents of a data signal can be suppressed.

It is another object of the present invention to provide an M-PSKmodulation method by which the degree of multiplexing can be increased.

SUMMARY OF THE INVENTION

To achieve the foregoing object and in accordance with an aspect of thepresent invention, there is provided a signal processing circuit whichcomprises a frequency conversion circuit for converting a spreadquadrature modulation signal of a received carrier band, which isobtained by performing a direct sequence type of a spectrum spreading onan in-phase component and a quadrature component of an informationsignal by using different spreading codes in a transmitting unit, into abaseband signal which has an in-phase component and a quadraturecomponent separated from each other, a first correlating detector for ade-spreading, which detects a correlation between the in-phase componentof the baseband signal and the spreading code corresponding to thein-phase component of the information signal, a second correlatingdetector for a de-spreading, which detects a correlation between thequadrature component of the baseband signal and the spreading codecorresponding to the in-phase component of the information signal, athird correlating detector for a de-spreading, which detects acorrelation between the in-phase component of the baseband signal andthe spreading code corresponding to the quadrature component of theinformation signal, a fourth correlating detector for a de-spreading,which detects a correlation between the quadrature component of thebaseband signal and the spreading code corresponding to the quadraturecomponent of the information signal, a decoding circuit for decodingdata from outputs of the first, second, third and fourth correlatingdetectors, a first multiplier for calculating a square of the output ofthe first correlating detector, a second multiplier for calculating asquare of the output of the second correlating detector, a first adderfor adding up outputs of the first and second multipliers, a judgmentcircuit for performing a synchronization acquisition judgement accordingto an output level of the first adder, a second adder for adding theoutputs of the first and second correlating detectors up, or adding theoutputs of the third and fourth correlating detectors up, or adding theoutputs of the first and fourth correlating detectors up, a subtracterfor performing a subtraction between the outputs of the second and thirdcorrelating detectors and a correction circuit for obtaining a frequencyoffset according to an output of the second adder and/or an output ofthe subtracter and performing a correction when the data is decoded.

Thus, in the receiving unit of the signal processing circuit of thepresent invention with the above described configuration, a quadraturedetection is first performed on a QPSK modulation wave or signal tosplit the QPSK modulation signal into an in-phase component and aquadrature component. Then, correlation detection is performed on one ofor both of the in-phase and quadrature components by using the spreadingcode C_(I) corresponding to the in-phase component (or channel) Ich ofthe transmitting unit and the spreading code C_(Q) corresponding to thequadrature component (or channel) Qch. Subsequently, synchronizationacquisition judgment is effected by using the sum of the squares ofcorrelation outputs obtained as results of the detection performed onthe same component by using the spreading codes C_(I) and C_(Q).Thereby, the influence of a frequency offset is eliminated.

Further, after the synchronization acquisition, the frequency offset isfound from the correlation outputs obtained by using the spreading codesC_(I) and C_(Q). Then, an AFC operation is performed.

Hereinafter, the operation of the circuit of the present invention willbe described by using equations practically. In case where thecorrelation detection is performed on the signals Ich and Qch expressedby the equations (9) and (10), respectively, by using the spreadingcodes C_(I) ' and C_(Q) ', the cross correlation C_(I) ×C_(Q) obtainedat the time of detecting the autocorrelations of the spreading codesC_(I) and C_(Q) is sufficiently small and can be neglected. Further,noises are averaged in one period of the spreading code. Thus, when thephase due to the frequency offset can be regarded as substantiallyconstant during one period of the spreading code, outputs of thecorrelators are obtained as follows.

    Σ{r(Ich)×c.sub.Ii '} MDI.sub.n cos Δωt(13)

    Σ{r(Ich)×c.sub.Qi '} -MDQ.sub.n sin Δωt(14)

    Σ{r(Qch)×c.sub.Ii '} MDI.sub.n sin Δωt(15)

    Σ{r(Qch)×c.sub.Qi '} MDQ.sub.n cos Δωt(16)

Then, the squares of the right sides of the equations (13) and (15) areadded together as follows. ##EQU2##

Thus, similarly as in case of employing a BPSK modulation, a synchronousdetection can be effected without being affected by the frequency offsetif the phase due to the frequency offset can be regarded assubstantially constant for one period of the spreading code.

Moreover, in case of the circuit of the present invention, the magnitudeof the frequency offset can be found by using the outputs of thecorrelators, which are expressed by the equations (13) to (16). Namely,tan Δωt can be obtained by using, for instance, the equations (13) and(15) as follows:

    MDI.sub.n sin Δωt/MDI.sub.n cos Δωt=tan Δωt                                           (18)

Thus, if -(π/2)≦Δωt≦(π/2), the value of Δωt can be obtained from thefollowing equation:

    Δωt=arc tan {MDI.sub.n sin Δωt/MDI.sub.n cos Δωt}                                          (19)

Incidentally, Δωt can be similarly obtained if the equations (14) and(16) are used instead of the equations (13) and (15).

Furthermore, as will be described hereunder, Δωt can be similarlyobtained if the equations (13) to (16) are used.

First, the following equation is obtained from the equations (13) and(16).

    MDI.sub.n cos Δωt+MDQ.sub.n cos Δωt=M(DI.sub.n +DQ.sub.n) cos Δωt                            (20)

Then, the following equation is obtained by subtracting the right sideof the equations (14) from that of the equation (15).

    MDI.sub.n sin Δωt+MDQ.sub.n sin Δωt=M(DI.sub.n +DQ.sub.n) sin Δωt                            (21)

Thus, tanΔωt can be obtained by using the equations (20) and (21) asfollows:

    M(DI.sub.n +DQ.sub.n) sin Δωt/M(DI.sub.n +DQ.sub.n) cos Δωt=tan Δωt                       (22)

Therefore, as is apparent from the equation (22), the value of Δωt canbe obtained similarly.

Additionally, an asynchronous quadrature detection is effected in thereceiving unit. Thus, a delay (or differential) detection can beperformed as follows. Namely, the AFC circuit can obtain informationconcerning Δω and then send the obtained information to the datadecoding circuit. Moreover, the outputs of the correlators expressed bythe equations (13) and (16) may be divided by (cosΔωt) and the frequencyoffset can be corrected. Further, an absolute synchronous detection canbe achieved by changing the frequency of a local signal, which is usedat the time of the modulation of the carrier band signal by using thebaseband signal, from exp(jωct) to exp(j(ωc-Δω)t) by use of Δω obtainedby the AFC circuit.

As described above, the synchronization acquisition can be effectedwithout being influenced by the frequency offset in case of the circuitof the present invention with the above-mentioned configuration.Therefore, when performing the method of sweeping the frequency of thecarrier within the maximum frequency offset range, a time required forthe synchronization acquisition can be reduced.

Further, in case of the circuit of the present invention, the magnitudeof the frequency offset can be found by using the outputs of thecorrelators. Thus the circuit of the present invention can be used notonly for a synchronous detection method, which needs a carrierregeneration, but also for a delay detection method in which thefrequency offset is corrected within the baseband immediately after theasynchronous quadrature detection.

Further, in accordance with another aspect of the present invention,there is provided a digital correlator which controls by using aspreading code signal whether or not the polarity of an input signal ora baseband reception signal obtained as a result of an A/D conversion isinverted. Alternatively, this digital correlator performs a processing,which is equivalent to such a control operation, on the input signal orthe reception signal. Then, in this correlator, an integrator integratesthe input signal for a predetermined period of time. Thereafter, thesquare of an output of the integrator is calculated. Further, in case ofemploying a modulation method by which signals are mapped onto a phaseplane (namely, a complex plane), the correlations corresponding to theaxes, respectively, are obtained. Then, the obtained correlations areadded together.

Thus, the correlator of the present invention with the above describedconfiguration employs a integrating method of the type that everysampling (namely, every A/D conversion), an addition or a subtraction isperformed on signals only one time. The correlator with a simpleconfiguration can be realized. Thereby, increase in operating frequency,as well as reduction in power consumption and in size of the circuit,can be achieved. Further, the correlator of the present invention, whichperforms a sampling a plurality of times for a time corresponding to 1bit of the spreading code, can obtain desired results.

Moreover, in accordance with a further aspect of the present invention,there is provided an M-PSK modulation method by which a differentialBPSK (hereunder abbreviated as DBPSK) modulation or a BPSK modulation isperformed on signals representing sequences of data to be codedcorrespondingly to symbol signals I and Q (namely, an in-phase componentand a quadrature component), by using the frequencies of carriers, thephases of which are deiffent from each other, in a transmitting unit anda resultant synthesized signal is transmitted therefrom, wherein a BPSKdelay detection or a BPSK detection is performed in an receiving unit onsignals obtained as results of quadrature detection by using outputsignals of corresponding correlators. Furthermore, in an embodiment, anangle of "phase rotation" (namely, phase shift) due to a frequencyoffset is estimated from outputs of correlators and is compensated andfurther, a decoding is effected by using values of correlationscorresponding to the components (or channels) I and Q, for the purposeof coping with the rotation on a phase plane.

Thus, in accordance with the present invention, even in case of theM-PSK modulation, a decoding of a baseband signal can be achieved byperforming a BPSK type decoding in a receiving unit of a spread spectrumcommunication system of the DS. As the result, the configuration of thecircuit of the receiving unit can be simplified. Moreover, theprocessing speed can be decreased. Further, when reproducing originaldata, data obtained as the result of the BPSK type decoding is decodedagain by using the employed code. Therefore, the larger M becomes, thegreater effects of the simplification become. Furthermore, even in caseof employing a BPSK delay detection, a data decoding with highreliability can be achieved under conditions (to be described later)similarly as in case of employing a synchronous detection of the M-PSKdemodulation system. Additionally, the frequency offset can becompensated by finding an angle of "phase rotation" due to the frequencyoffset from outputs of corresponding correlators and correcting thefrequency of an oscillator of the receiving unit, or by cancelling the"phase rotation" of data when decoding the data and making a judgment onthe data. Here, an output of the correlator corresponding to a spreadingcode (i), which corresponds to the in-phase component, and to thecomponent I of the receiving unit becomes 0 due to the "(phase)rotation" on the phase plane when the angle of "phase rotation" is equalto (π/2) or π. Thus, by compensating the frequency offset by adding upoutputs of correlators respectively corresponding to the components Iand Q, it becomes possible to omit a circuit for distinguishing a casewhere such an addition is not performed. Consequently, the constructionof the circuit can be further simpler. Further, effects of thesimplification of the configuration of the circuit can be increased.Furthermore, the characteristics of the system employing a multiplexingcan be improved.

Furthermore, in accordance with yet another aspect of the presentinvention, there is provided a spreading code generating method in whichcodes each having a good autocorrelation are employed as differentspreading code patterns used correspondingly to phase components,respectively, and the spreading code patterns are multiplied by a sameorthogonal code having good cross correlation characteristics togenerate a spreading code pattern. Thereby, the degree of multiplexingcan be increased.

Thus, the spreading code pattern can be generated by utilizing thecharacteristics of the codes. (In an embodiment, a PN code is used asthe codes having a good autocorrelation.) Further, the interferencebetween spread signals can be decreased. Thus, for instance, a QPSKsignal can be split to the components Ich and Qch by using a PN code.Namely, one orthogonal code can be assigned to a QPSK modulation signal.

At that time, if an intersymbol synchronization is taken, there is nocross correlation between signals respectively having phase components,to which a same PN code pattern is assigned, due to the orthogonal code.Thus, the interference can be reduced and the degree of multiplexing canbe increased.

In contrast, even if no intersymbol synchronization is taken, the phasecomponents are multiplied by the PN code. Thus the deterioration of thecharacteristics of the cross correlation is reduced in comparison withthe case where only the orthogonal code is used, and the characteristicsof the cross correlation are almost the same as those of theconventional case employing a PN code. Similarly, the problem of theprior art, namely, the difficulties in effecting the synchronizationacquisition and the multipath separation can be overcome due to the highautocorrelation of the PN code.

However, there is restriction on the combination of code patterns of agiven spreading code, which have a small cross correlation. Meanwhile,when causing a phase offset in code patterns having large crosscorrelations, in a circulating manner, the cross correlations vary andsometimes the code patterns have a minimum value of the crosscorrelation therebetween at a given phase offset. Thus, anotherspreading code generating method is devised according to the presentinvention, in view of the fact that an intersymbol synchronization isalways established in cases of the phase components of given M-PSKmodulation signals.

Namely, in accordance with an additional aspect of the presentinvention, there is provided a spreading code generating method in whichamong spreading code patterns comprised of spreading codes used forM-PSK modulation of the DS type, spreading code patterns each having amaximum value of the cross correlation larger than the values of thecross correlations of the other spreading code patterns are employed asthe different spreading code patterns used correspondingly to the phasecomponents, and circulating the employed spreading code patternsmutually and causing a phase offset therein at a point where the crosscorrelation between the codes of the employed spreading code patternshas a minimum value.

Thereby, the interference between the spread waves can be suppressed andsimultaneously, the cross correlation between the patterns can beminimized when the phase components are not completely separated fromeach other in the receiving unit. This utilizes the fact that the phasecomponents are always synchronized with each other and the value of thecross correlation between the phase components is constant, as describedabove. Additionally, if another spread wave or signal interferes withone of the phase components, there is no correlation between this spreadwave and the other phase component. Consequently, the possibility of anoccurrence of malfunction in a synchronization acquisition andmaintenance or the like is reduced.

BRIEF DESCRIPTION OF THE DRAWINGS

Other features, objects and advantages of the present invention willbecome apparent from the following description of preferred embodimentswith reference to the drawings in which like reference charactersdesignate like or corresponding parts throughout several views, and inwhich:

FIG. 1 is a schematic block diagram for illustrating the configurationof a conventional synchronization acquisition circuit of BPSK type;

FIG. 2(a) is a schematic block diagram for illustrating theconfiguration of a spreading circuit of a conventional spread spectrumcommunication system;

FIG. 2(b) is a schematic block diagram for illustrating theconfiguration of a de-spreading circuit of the conventional spreadspectrum system of FIG. 2(a);

FIG. 3 is a detail block diagram for illustrating the configuration of aconventional correlator which employs a digital matched filterpractically;

FIG. 4(a) is a schematic block diagram for illustrating the transmittingunit of a conventional spread spectrum communication system of thedirect sequence type that employs a QPSK;

FIG. 4(b) is a schematic block diagram for illustrating the receivingunit of the conventional spread spectrum communication system of FIG.4(a);

FIG. 5 is a schematic block diagram for illustrating the configurationof a primary part of a signal processing circuit embodying the presentinvention (namely, a first embodiment of the present invention);

FIG. 6 is a schematic block diagram for illustrating the configurationof a primary part of another signal processing circuit embodying thepresent invention (namely, a second embodiment of the presentinvention);

FIG. 7 is a schematic block diagram for illustrating the configurationof a digital correlator embodying the present invention (namely, a thirdembodiment of the present invention);

FIGS. 8(a) and 8(b) are timing charts each for illustrating an operationof the third embodiment of the present invention;

FIG. 9 is a schematic block diagram for illustrating the configurationof another digital correlator embodying the present invention (namely, afourth embodiment of the present invention);

FIG. 10 is a timing chart for illustrating an operation of the thirdembodiment of the present invention;

FIG. 11 is a schematic block diagram for illustrating the configurationof a further digital correlator embodying the present invention (namely,a fifth embodiment of the present invention);

FIG. 12 is a schematic block diagram for illustrating the configurationof still another digital correlator embodying the present invention(namely, a sixth embodiment of the present invention);

FIG. 13 is a schematic block diagram for illustrating the configurationof yet another digital correlator embodying the present invention(namely, a seventh embodiment of the present invention);

FIG. 14 is a schematic block diagram for illustrating the configurationof a part of a receiving circuit of a spread spectrum communicationsystem of the DS type employing a QPSK, which is provided with digitalcorrelators of the present invention (namely, an eighth embodiment ofthe present invention);

FIG. 15(a) is a schematic block diagram for illustrating theconfiguration of a transmitting circuit of a spread spectrumcommunication system of the DS type according to an M-PSK modulationmethod of the present invention, which is a ninth embodiment of thepresent invention;

FIG. 15(b) is a schematic block diagram for illustrating theconfiguration of a receiving circuit of the spread spectrumcommunication system of the DS type of FIG. 15(a);

FIG. 16(a) is a schematic block diagram for illustrating theconfiguration of a receiving circuit of a spread spectrum communicationsystem of the DS type according to another M-PSK modulation method ofthe present invention, which is a tenth embodiment of the presentinvention;

FIG. 16(b) is a schematic block diagram for illustrating theconfiguration of a modification of the receiving circuit of the spreadspectrum communication system of the DS type of FIG. 16(a);

FIG. 17(a) is a schematic block diagram for illustrating theconfiguration of a transmitting circuit of a spread spectrumcommunication system of the DS type according to a further M-PSKmodulation method of the present invention, which is an eleventhembodiment of the present invention;

FIG. 17(b) is a schematic block diagram for illustrating theconfiguration of a receiving circuit of the spread spectrumcommunication system of the DS type of FIG. 17(a);

FIG. 18(a) is a diagram for illustrating a method for compensating afrequency offset in an M-PSK modulation system of the present invention;

FIG. 18(b) is a schematic block diagram for illustrating theconfiguration of a receiving circuit of a spread spectrum communicationsystem of the DS type according to still another M-PSK modulation methodof the present invention, which is a twelfth embodiment of the presentinvention;

FIG. 19(a) is a graph for showing C/N-BER characteristics (namely, therelation between the carrier-to-noise-ratio and the bit error rate) ofthe eleventh embodiment of the present invention;

FIG. 19(b) is a graph for showing frequency-offset characteristics ofthe eleventh embodiment of the present invention;

FIG. 20(a) is a schematic block diagram for illustrating theconfiguration of a transmitting unit of a DS/DPSK system for performinga spreading code generating method according to the present invention(namely, a thirteenth embodiment of the present invention);

FIG. 20(b) is a schematic block diagram for illustrating theconfiguration of a receiving unit of the DS/DPSK system of FIG. 20(a);

FIG. 21 is a diagram for illustrating the scheme of the DS/DPSK systemof FIG. 20(a);

FIG. 22 is a timing chart for illustrating the relation among PN codesignals and a Hadamard Function (hereunder abbreviated as HF) signal;

FIG. 23 is a diagram for illustrating patterns of codes employed in aDS/QPSK system for performing a spreading code generating methodaccording to the present invention (namely, a fourteenth embodiment ofthe present invention);

FIG. 24 is a diagram for illustrating practical examples of quadraturespreading code sets generated in the fourteenth embodiment of thepresent invention; and

FIG. 25 is a diagram for illustrating maximum values of the crosscorrelations between the code patterns of FIG. 23.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Hereinafter, the preferred embodiments of the present invention will bedescribed in detail by referring to the accompanying drawings.

(Embodiment 1)

First, a first embodiment of the present invention will be describedhereinbelow in detail by referring to FIG. 5.

FIG. 5 is a schematic block diagram for illustrating the configurationof a primary part (namely, synchronization acquisition and AFC circuits)of a signal processing circuit embodying the present invention (namely,the first embodiment of the present invention).

In this figure, reference numeral 10 designates a radio antenna; 20 aBPF for transmitting a definite band of frequencies of a signal receivedfrom the radio antenna 10; 30 an AGC circuit for performing an automaticgain control operation on an output of the BPF 20; 41 a down-mixercorresponding to the in-phase component (or channel) Ich of the signal;42 a down-mixer corresponding to the quadrature component (or channel)Qch of the signal; 51 a local signal generator; 52 a (π/2)-phaseshifter; 61 and 62 LPFs; 71 and 72 A/D converters; 81 to 84 correlatingdetectors each for a de-spreading; 91 and 92 squaring devices each forcalculating the square of an input signal; 600 an AFC circuit; 610 adata decoding circuit; 651 an adder for adding up outputs of thecorrelators 81 and 84; 652 an adder for adding up an output of thecorrelator 82 and a negative output of the correlator 84; 620 an adderfor adding up outputs of the squaring device 91 and 92; 630 asynchronization acquisition judgment circuit; 640 a spreading codegenerating circuit; and 700 a code clock recovery circuit.

When the radio antenna 10 receives a spread quadrature modulation waveor signal having an in-phase component Ich and a quadrature componentQch which have undergone a spectrum spreading process using a spreadingcode C_(I) and another spectrum spreading process using a spreading codeC_(Q), respectively (incidentally, in each of these processes, what iscalled the spreading code rate is N times per symbol), the spreadquadrature modulation signal received by the antenna 10 is sent throughthe BPF 20 and the ACC circuit 30 to the mixers 41 and 42 for performinga frequency conversion and converting a signal inputted thereto into abaseband signal. Then, in the mixers 41 and 42, the spread quadraturemodulation signal (namely, the reception signal) is down-converted intoan in-phase component Ich and a quadrature component Qch which havefrequencies of the baseband, by using carriers which have been generatedby the local signal generator 51 and inputted to the mixer 41 and to themixer 42 through the phase-shifter 52 for performing a quadratureseparation.

The components Ich and Qch of the baseband signal converted from thereception signal is converted at the spreading code rate by the A/Dconverters 71 and 72 into digital signals after passing through theLPFs. Simultaneously with this, the signal Ich and Qch having passedthrough the LPFs are sent to the code clock recovery circuit Thereupon aspreading code clock is recovered.

After the A/D conversion, correlation detections are performed on thesignals Ich and Qch each data symbol interval in the correlators byusing the spreading codes C_(I) and C_(Q) generated by the spreadingcode generator 640. Namely, the correlators (namely, the correlatingdetectors) 81, 82, 83 and 84 detect the correlation between the signalIch represented by the equation (13) and the spreading code C_(I), thatbetween the signal Ich represented by the equation (14) and thespreading code C_(Q), that between the signal Qch represented by theequation (15) and the spreading code C_(I), and that between the signalQch represented by the equation (16) and the spreading code C_(Q),respectively. Further, a synchronization acquisition detection isperformed according to the equation (17) by using a value obtained bythe adder 620. Namely, the adder 620 adds up outputs of the squaringdevices 91 and 92 which calculate the square of a value of thecorrelation detected by the correlator 81 and that of a value of thecorrelation detected by the correlator 83, respectively. Then, thesynchronization acquisition judgment circuit 630 judges from the outputof the adder 620 whether or not the synchronization acquisition isachieved. Further, an initial synchronization acquisition is effected bycontrolling the patterns of codes generated by the spreading codegenerating circuit 640.

Next, when the synchronization acquisition of the spreading code isestablished, the AFC circuit 600 starts a frequency offset correctionprocessing. In case of this embodiment, the frequency offset is obtainedaccording to the equation (22). Thus, the frequency offset Δω isobtained in the AFC circuit 600 by using an output of the adder 651(namely, a result of adding up correlation outputs of the correlators 81and 84) and an output of the adder 652 (namely, a result of adding theinverse of a correlation output of the correlator 82 to a correlationoutput of the correlator 84). Then, correction value information is sentto the data decoding circuit 610 therefrom. Subsequently, the datadecoding circuit 610 decodes data according to the outputs of thecorrelators 81 to 84 and the AFC circuit 600.

As described above, this embodiment is provided with the correlators fora de-spreading, which correspond to the different combinations of thespreading codes C_(I) and C_(Q) and the components (or channels) Ich andQch obtained as the result of the quadrature detection. Further, thesynchronization acquisition judgment is effected by using a sum of thesquares of outputs of the correlators. Thus, in case where the phase dueto the frequency offset corresponding to one period of the spreadingcode can be regarded as almost constant, the influence of the frequencyoffset can be eliminated. Moreover, a time required for asynchronization acquisition when effecting an initial synchronizationacquisition can be reduced. Furthermore, the frequency offset can beobtained by using outputs of the correlators. Consequently, whendecoding data, the frequency offset correction can be realized.

Incidentally, the circuit of this embodiment is constructed to obtainthe frequency offset according to the equation (22). However, thecircuit of this embodiment may be modified to obtain the frequencyoffset according to the equation (19). In such a case, the adders 651and 652 are removed from the configuration of FIG. 1. Moreover, thecombination of outputs of the correlators 81 and 83 is used for thesynchronization acquisition judgement. However, if the combination ofoutputs of the correlators 82 and 84 is used, a synchronizationacquisition judgment can be similarly performed. Furthermore, in case ofthis embodiment, outputs of the four correlators are used for decodingdata. However, the combination of outputs of the correlators 82 and 83may be used for decoding data, instead of the outputs of the fourcorrelators.

(Embodiment 2)

Next, a second embodiment of the present invention will be describedhereinbelow in detail by referring to FIG. 6.

FIG. 6 is a schematic block diagram for illustrating the configurationof a primary part (namely, synchronization acquisition and AFC circuits)of a signal processing circuit embodying the present invention (namely,the second embodiment of the present invention).

In this figure, reference numeral 10 designates a radio antenna; 20 aBPF; 30 an AGC circuit; 41 a down-mixer corresponding to the in-phasecomponent (or channel) Ich of the signal; 42 a down-mixer correspondingto the quadrature component (or channel) Qch of the signal; 51 a localsignal generator; 52 a (π/2)-phase shifter; 61 and 62 LPFs; 71 and 72A/D converters; 81, 83 and 84 correlating detectors (namely,correlators) each for a de-spreading; 91 and 92 squaring devices; 600 anAFC circuit; 610 a data decoding circuit; 620 an adder; 630 asynchronization acquisition judgment circuit; 640 a spreading codegenerating circuit; and 700 a code clock recovery circuit. Thesecomposing elements of the second embodiment of the present invention aresimilar to the corresponding elements of the first embodiment of FIG. 5.The second embodiment of FIG. 6 is different from the first embodimentof FIG. 5 in that regarding the signal Qch, the correlation between thissignal and each of the spreading codes C_(I) and C_(Q) is obtained,while only the correlation between the spreading code C_(I) and thesignal Ich is obtained, and that the correlating detector 82 and theadders 651 and 652 are removed from the configuration of FIG. 5.

Further, a synchronization acquisition judgment operation of the secondembodiment of the present invention is similar to that of the firstembodiment of the present invention. However, the calculation of thefrequency offset to be effected upon completion of the synchronizationacquisition is performed according to the equation (19). Thus, the AFCcircuit obtains the value of the frequency offset Δω from outputs of thecorrelators 81 and 83 and then sends the obtained value to the datadecoding circuit 610 as correction value information. Subsequently, thedata decoding circuit decodes data according to the outputs of thecorrelators 81 and 84 and the correction value information sent from theAFC circuit 610.

However, as the result of obtaining the correlation between the signalQch and each of the spreading codes C_(I) and C_(Q) and the correlationbetween the spreading code C_(I) and the signal Ich after the quadraturedetection, and then calculating the frequency offset Δω from the outputsof the correlators, effects similar to those of the first embodiment canbe obtained. Namely, the influence of the frequency offset can beeliminated. Moreover, the configuration of the circuit can besimplified.

Incidentally, in case of the second embodiment, the correlation betweenthe signal Qch and each of the spreading codes C_(I) and C_(Q) and thecorrelation between the spreading code C_(I) and the signal Ich areobtained. However, instead of this, the correlation between the signalIch and each of the spreading codes C_(I) and C_(Q) and the correlationbetween the spreading code C_(Q) and the signal Qch may be obtained.

In addition, as can be understood from the facts that each of the firstand second embodiments of the present invention employs the spreadingcode generating circuit, it is assumed that an active correlator is usedas the correlating detector for a de-spreading. However, instead of sucha correlator, another type correlator such as a matched filter may beused. Moreover, in cases of the first and second embodiments of thepresent invention, the A/D conversion is performed at the spreading coderate. However, the A/D conversion may be effected at another rate higherthan the code spreading rate.

Furthermore, in cases of the first and second embodiments of the presentinvention, after the asynchronous quadrature detection, the frequencyoffset correction is effected by the AFC circuit when decoding data.However, because the value of the frequency offset Δω is obtained asdescribed above, AFC operation can be performed according to what iscalled a carrier recovery method in an absolute synchronizationdetection circuit.

Hereinafter, the third to eighth embodiments of the present inventionwill be described. Incidentally, in the descriptions of the third toeighth embodiments of the present invention, spreading code signals areindicated by "+1" and "-1". Further, when performing an A/D conversion,the center level of an input analog signal is assumed to be "0" and theanalog input signal is converted into a "+" component (or digitalsignal) and a "-" component (or digital signal).

(Embodiment 3)

Hereunder, the third embodiment of the present invention will bedescribed by referring to FIGS. 7, 8(a) and 8(b).

FIG. 7 illustrates the configuration of a digital correlator embodyingthe present invention (namely, the third embodiment of the presentinvention). Further, FIGS. 8(a) and 8(b) are timing charts forillustrating operations of the digital correlator of FIG. 7.

In FIG. 7, reference numeral 711 denotes an amplifier, the inverting ornon-inverting of an input signal to which is controlled according to aspreading code signal; 712 an A/D converter; 713 an integrator foradding up outputs of the A/D converter 712 for a preliminarily setperiod of time; and 714 a squaring device for obtaining the square of anoutput of the integrator 713.

Hereinafter, operations of the digital correlator (namely, the thirdembodiment of the present invention) having the configuration describedhereinabove will be described by referring to FIGS. 8(a) and 8(b).Further, FIG. 8(a) is a timing chart for illustrating an operation ofthis digital correlator in case where there is no correlation between aspreading code signal and an input signal inputted thereto. Moreover,FIG. 8(b) is a timing chart for illustrating an operation of thisdigital correlator in case where there is a correlation between aspreading code signal and an input baseband signal inputted thereto.

In FIG. 8(a), reference characters S₁ and S₂ designate an input basebandsignal and a spreading code signal inputted to the amplifier 711. If thepolarity of the spreading code signal is positive (+), an input signalof the amplifier 711 is not inverted. In contrast, if the polarity ofthe spreading code signal is negative (-), an input signal of theamplifier 711 is inverted as indicated by S₃ in this figure. Then, anA/D conversion is performed on such an input signal and as aconsequence, a digital signal is obtained as indicated by S₄.Subsequently, the integrator 713 integrates this digital signal S₄ for apredetermined period of time (for example, one (data) symbol interval orone period of the spreading code signal). In case that there is acorrelation between the digital (input) signal and the spreading codesignal, an output of the integrator increases gradually in such apredetermined period of time, as indicated by a curve S₅. In contrast,in case where there is no correlation (or synchronism) between an inputsignal and a spreading code signal, as indicated by S₆ and S₇ in FIG.8(b), the waveform of the input signal inverted according to thepolarity of the spreading code signal becomes like a curve indicated byreference character S₈ in this figure. Further, as a result of an A/Dconversion of such an input signal, a digital signal indicated by S₉ isobtained. Moreover, a signal representing the result of an integrationof this digital signal becomes as indicated by S₁₀. Thus, in this case,an output of the integrator is not large. Incidentally, this figureillustrates a case that the polarity of a data signal (namely, the inputsignal) is positive (+). In contrast, in case where the polarity of thedata signal is negative (-), an output of the integrator becomesnegative. Thus, the correlation between the data signal (namely, theinput signal) and the spreading code signal is obtained by calculatingthe square of an output of this integrator 713 by use of the squaringdevice 714. Incidentally, in case that there is a correlation betweenthe data signal and the spreading code signal, an output of theintegrator 713 can be used for decoding data. Thereby, a digitalcorrelator with a simple configuration can be realized by performing theprocess described hereinbefore.

(Embodiment 4)

Hereinafter, the fourth embodiment of the present invention will bedescribed by referring to FIGS. 9 and 10.

In FIG. 9, reference numerals 712, 713 and 714 designate an A/Dconverter, an integrator and a squaring device, respectively, similarlyas in case of the third embodiment of the present invention. Further,reference numeral 715 denotes a signal processing circuit for invertingthe polarity of an output of the A/D converter 712 according to thepolarity (or sign) of the spreading code signal.

In case of this embodiment having such a configuration, an A/Dconversion is performed on an analog input signal S₁₁ directly. As theresult of the A/D conversion, a digital signal as indicated by S₁₃ inFIG. 10 is obtained. Then, an inverting or non-inverting of such anoutput signal of the A/D converter 712 is effected in the signalprocessing circuit 715 according to the polarity of the spreading codesignal S₁₂. Therefore, an output signal of the signal processing circuit715 becomes equivalent to a digital signal S₁₄ which would be obtainedif an A/D conversion from the analog input signal to such a digitalsignal were performed. Subsequently, the integrator 713 integrates thisdigital signal and further the square of an output signal of theintegrator 713 is obtained by the squaring device 714 to find acorrelation value.

As described above, an inverting or non-inverting processing of ananalog signal is not performed. Thus, the fourth embodiment of thepresent invention is advantageous to a conversion from an analog inputsignal to a digital signal.

(Embodiment 5)

Next, the fifth embodiment of the present invention will be describedhereinbelow by referring to FIG. 11. As shown in this figure, thisembodiment has two systems each consisting of an A/D converter 712 and aswitch 716 which is constructed by a register practically. Further, theselection of one of signals respectively passing through these systemsis according to the polarity of a spreading code signal. When thepolarity of the spreading code signal is positive (+), the switch 716ais turned on and then the integrator 713 integrates a signal obtained asa result of an A/D conversion performed by the A/D converter 712a on aninput signal which is not inverted. In contrast, when the polarity ofthe spreading code signal is negative (-), the switch 716b is turned onand then the integrator 713 integrates a signal obtained as a result ofan A/D conversion performed by the A/D converter 712b on an invertedinput signal sent thereto through an inverting amplifier 711, the gainof which is 1. Finally, a correlation value is obtained by calculatingthe square of an output of the integrator 713 by the squaring device714. The correlation value obtained by this embodiment in this way issimilar to that obtained by the fourth embodiment. However, in case ofthe fifth embodiment of the present invention, an inverting ornon-inverting of an input signal is controlled by utilizing theturning-on and turning-off of the switches 716a and 716b instead ofusing an amplifier as in case of the third embodiment which uses theamplifier 711. Thus, timing operations in the correlator are simplified.

(Embodiment 6)

Next, the sixth embodiment of the present invention will be describedhereinbelow by referring to FIG. 12. As shown in this figure, twointegrators 713a and 713b and an adder 717 are provided in thisembodiment. Further, one of the integrators 713a and 713b are selectedaccording to the polarity of a spreading code signal. Namely, when thepolarity of the spreading code signal is positive (+), the integrator713a is used to integrate a digital signal obtained by performing an A/Dconversion of an analog input signal. In contrast, when the polarity ofthe spreading code signal is negative (-), the integrator 713b is usedto integrate the digital signal. In this case, the adder 717 inverts anoutput of the integrator 713b. Finally, the squaring device 714calculates the square of an output of the adder 717 to obtain acorrelation value. In case of this embodiment, an operation of invertingthe digital signal is carried out only once in each rest time. Thus,similarly as in case of the fifth embodiment, timing operations can beeffected in the correlator easily.

(Embodiment 7)

Next, the seventh embodiment of the present invention will be describedhereinbelow by referring to FIG. 13. As is seen from this figure, anintegrator 719 of the type being capable of performing addition andsubtraction operations is employed in this embodiment in place of thetype of the integrator 713a or 713b of the sixth embodiment. Thus, theintegrator 719 integrates an input digital signal by changing anoperation between an addition and a subtraction according to thepolarity of the spreading code signal. Then, the squaring device 714calculates the square of an output of the integrator 719 to obtain acorrelation value. Although the seventh embodiment of the presentinvention requires the integrator 719 of the type being capable ofperforming addition and subtraction operations, the configuration ofthis embodiment is very simple.

(Embodiment 8)

Next, the eighth embodiment of the present invention will be describedhereinbelow by referring to FIG. 14. This figure illustrates a part of areceiving circuit of the spread spectrum communication system of the DStype employing a modulation method such as a QPSK modulation method bywhich input signals are mapped onto a phase plane. In this figure,reference numerals 781_(i) and 781_(q) designate frequency converterscorresponding to the in-phase component I and the quadrature componentQ, respectively; 783 a local oscillator; 784_(i) and 784_(q) LPFscorresponding to the in-phase component I and the quadrature componentQ, respectively; 785_(i) a digital correlator for obtaining acorrelation between the in-phase component I of a signal transmittedfrom a transmission unit and the spreading code signal corresponding tothe in-phase component I of a signal received by a receiving unit;785_(q) a digital correlator for obtaining a correlation between thein-phase component I of a signal transmitted from a transmission unitand the spreading code signal corresponding to the quadrature componentQ of a signal received by a receiving unit; 786_(i) a digital correlatorfor obtaining a correlation between the quadrature component Q of asignal transmitted from a transmission unit and the spreading codesignal corresponding to the in-phase component I of a signal received bya receiving unit; 786_(q) a digital correlator for obtaining acorrelation between the quadrature component Q of a signal transmittedfrom a transmission unit and the spreading code signal corresponding tothe quadrature component Q of a signal received by a receiving unit;787_(i) a squaring device for obtaining the square of the value of thecorrelation between the in-phase component I of a signal transmittedfrom a transmission unit and the spreading code signal corresponding tothe in-phase component I of a signal received by a receiving unit;787_(q) a squaring device for obtaining the square of the value of thecorrelation between the in-phase component I of a signal transmittedfrom a transmission unit and the spreading code signal corresponding tothe quadrature component Q of a signal received by a receiving unit; 782a phase shifter for shifting the phase of a signal sent from the localoscillator 783 by (π/2); and 788 an adder.

In the circuit having the configuration described hereinabove, a signalreceived by a front-end portion of the receiving unit is first processedto increase the signal level thereof to a necessary level. Then, thereceived signal is inputted to the frequency converters 781_(i) and781_(q) which convert the inputted signal into baseband signalscorresponding to the components I and Q, respectively. Thereafter, thebaseband signal corresponding to the components I is inputted throughthe LPF 784_(i) to the digital correlators 785_(i) and 786_(i) whichemploy one of methods of the third to seventh embodiments. Further, thebaseband signal corresponding to the components Q is inputted throughthe LPF 784_(q) to the digital correlators 785_(q) and 786_(q) whichemploy one of methods of the third to seventh embodiments. Incidentally,for instance, the correlator 785i finds the correlation between thebaseband signal obtained by the frequency converter 781i and thespreading code signal corresponding to the component I of a signaltransmitted from the transmitting unit. Generally, it is often that thefrequency of a local oscillator of a transmitting unit (for example, incase of the conventional system, the local oscillator 904 of FIG. 2(a))is not precisely equal to that of a local oscillator of a receiving unit(for instance, in case of the conventional system, the local oscillator910 of FIG. 2(b)). Therefore, in case of the system of FIG. 14, on aphase plane, a signal output mapped thereto rotates around the originthereof. Thus in case that only one correlator is used correspondinglyto each of the components I and Q, outputs of the correlators fluctuateand can not be obtained stably. However, this problem can be solved byproviding two correlators, which correspond to the spreading codesignals corresponding to the in-phase and quadrature components I and Q,respectively, correspondingly to each of the baseband signals I and Qand then adding up the squares of outputs of the correlatorsrespectively obtaining the correlations between one I or Q of thecomponents of a signal transmitted from a transmission unit and thespreading code signal corresponding to the in-phase component I of asignal received by a receiving unit and the correlation between the sameone of the components of the signal transmitted from the transmissionunit and the spreading code signal corresponding to the in-phasecomponent Q of the signal received by the receiving unit. Thereby,outputs of the correlators can be obtained stably. In case of thecircuit of FIG. 14, a correlation output corresponding to the spreadingcode signal which corresponds to the in-phase component I of a signaltransmitted from the transmitting unit is obtained by using the adder786 for adding up the square of an output of the correlator 785i, whichis calculated by the squaring device 787i, and the square of an outputof the correlator 785q, which is calculated by the squaring device 787q.Further, note that similarly, an correlation output corresponding to thespreading code signal which corresponds to the quadrature component Q ofa signal transmitted from the transmitting unit is obtained by using theadder 788 for adding up the square of an output of the correlator 786iand the square of an output of the correlator 786q and that a syntheticcorrelation output can be obtained by using both of the correlationoutputs corresponding to the spreading code signals which correspond tothe components I and Q (for example, averaging these correlationoutputs). Moreover, any one of the third to seventh embodiments may beused as each of the correlators of the eighth embodiment. Thereby, theconfiguration of the circuit of the eighth embodiment of the presentinvention can be extremely simplified.

Incidentally, in the foregoing descriptions of the third to eighthembodiments of the present invention, the sampling number is 10 per oneperiod of the spreading code signal (as illustrated in FIGS. 8(a), 8(b)and 10). However, another sampling number (for instance, 1 or 2 per oneperiod of the spreading code signal) may be used to obtain similarcorrelation value.

(Embodiment 9)

Hereinafter, a ninth embodiment of the present invention will bedescribed hereinbelow by referring to the accompanying drawings.

FIG. 15(a) is a schematic block diagram for illustrating theconfiguration of a transmitting circuit (namely, a spreading portion) ofa spread spectrum communication system of the DS type according to anM-PSK modulation method of the present invention (namely, the ninthembodiment of the present invention). Further, FIG. 15(b) is a schematicblock diagram for illustrating the configuration of a receiving circuit(namely, a de-spreading portion) of the spread spectrum communicationsystem of the DS type of FIG. 15(a).

In FIG. 15(a), reference numeral 2101 denotes an encoder for convertingan input signal into data sequences corresponding to two symbols orcomponents I and Q, respectively; 2102i a differential encoder forperforming a differential encoding of the data sequence corresponding tothe symbol I, which is outputted by the encoder 2101; 2102q adifferential encoder for performing a differential encoding of the datasequence corresponding to the symbol Q, which is outputted by theencoder 2101; 2103i a multiplier (or mixer) for mixing an output signalof the differential encoder 2102i with a spreading code signal sent froma spreading code generator 2104i; 2103q a multiplier (or mixer) formixing an output signal of the differential encoder 2102q with aspreading code signal sent from a spreading code generator 2104q; 2109 aquadrature modulation portion comprised of a multiplier 2105i for mixingan output signal of the multiplier 2103i with a signal sent from a firstlocal oscillator 2107, a multiplier 2105q for mixing an output signal ofthe multiplier 2103q with a signal received through a(π/2)-phase-shifter 2106 from the first local oscillator 2107 and anadder 2108 for adding outputs of the multipliers 2105i and 2105q; 2110 amultiplier for mixing an output of the quadrature modulation portion2109 with a signal sent from a second local oscillator 2111; and 2112 anantenna.

In the circuit with the configuration of this figure, an input signal isfirst inputted to the encoder 2101 for mapping the input signal onto asymbol signal used for a QPSK modulation, whereupon the input signal isconverted into the data sequences corresponding to the two symbols I andQ. respectively. Then, these data sequences corresponding to I and Qundergo differential encoding operations in the differential encoders2102i and 2102q, respectively. Subsequently, an output of thedifferential encoder 2102i is multiplied by a corresponding spreadingcode signal, which is sent from the spreading code generator 2104i, inthe modulo-2 multiplier 2103i. Similarly, an output of the differentialencoder 2102q is multiplied by a corresponding spreading code signal,which is sent from the spreading code generator 2104q, in the modulo-2multiplier 2103q. Then, outputs of the multipliers 2103i and 2103q aresent to the quadrature modulation portion 2109. Further, in thequadrature modulation portion 2109, the output of the multiplier 2103iis mixed with a signal sent from the first local oscillator 2107 in themixer 2105i. Moreover, the output of the multiplier 2103q is mixed inthe mixer 2105q with a signal sent from the first local oscillator 2107,the phase of which is shifted by (π/2) by the (π/2)-phase shifter 2106.Incidentally, the modulation method employed at that time is the DBPSKmodulation to be performed on a quadrature phase plane. Then, outputs ofthe mixers 2105i and 2105q are added together by the adder 2108.Further, the spectrum of an output signal of the adder 2108 is similarto that of a signal obtained as a result of a QPSK modulation.Thereafter, the output signal of the adder 2108 is converted by themixer 2110 into a signal of a carrier band by using a signal sent fromthe second local oscillator 2111. Then, the signal of a carrier band istransmitted from the antenna 2112.

On the other hand, in FIG. 15(b), reference numeral 2113 designates anantenna for receiving the signal transmitted from the antenna 2112; 2114a multiplier for mixing an input signal, which is received from theantenna 2113 and should be modulated, with a signal sent from a firstlocal oscillator 2115; 2118 a quadrature detection portion consisting ofa multiplier 2116i for mixing an output of the multiplier 2114 with asignal sent from a second local oscillator 2119 and a multiplier 2116qfor mixing an output of the multiplier 2114 with a signal sent through a(π/2)-phase shifter 2117 from the second local oscillator 2119; 2120a to2120d correlators for obtaining the correlations between one of outputsof the multipliers 2116i and 2116q and each of the spreading codesignals 2121i and 2121q and between the other of outputs of themultipliers 2116i and 2116q and each of the spreading code signals 2121iand 2121q; 2122i and 2122q BPSK delay detection portions; and 2123 aQPSK decoder.

In the receiving circuit of FIG. 15(b) having the configurationdescribed hereinabove, first, a signal of an intermediate frequencyband, which is obtained by the mixer 2114, is converted by thequadrature detection portion 2118 into a signal of a baseband.Thereafter, data signals are obtained by using the four correlators2120a to 2120d. In case of this circuit, the BPSK delay detection isperformed by the BPSK delay detection circuit 2122i by using an outputV₁ of the correlator 2120a and an output V₃ of the correlator 2120c.Simultaneously, the BPSK delay detection is also performed by the BPSKdelay detection circuit 2122q by using an output V₂ of the correlator2120b and an output V₄ of the correlator 2120d. Thus, a signalrepresenting the data sequence corresponding to the component I andanother signal representing the data sequence corresponding to thecomponent Q are detected. At that time, outputs of the correlators varyas the "phase rotation" due to the frequency offset increases.Therefore, the detection is effected by changing the outputs of thecorrelators, for instance, between V₁ and V₃. Thereafter, transmissiondata is decoded by using the QPSK decoder 2123.

As described above, in case of this embodiment, the transmission data istransmitted and received similarly as in case of employing a QPSKmodulation. However, the detection method effected in this embodimentpractically is the quadrature multiplex DBPSK modulation. Thus, thisembodiment has an advantageous effect in that the system is resistant tothe influence of the frequency offset, in comparison with theconventional system employing the QPSK modulation.

(Embodiment 10)

Hereinafter, the tenth embodiment of the present invention will bedescribed by referring to FIGS. 16(a) and 16(b).

FIG. 16(a) illustrates the configuration of a receiving portion of aspread spectrum communication system of the DS type corresponding to theQPSK modulation similarly to that of the ninth embodiment of FIG. 15(a).The receiving portion of the tenth embodiment is different from that ofthe ninth embodiment in that the tenth embodiment is provided with afrequency offset compensation circuit.

Namely, the receiving circuit of the tenth embodiment of the presentinvention is provided with a phase detecting circuit 2124 for detectingan angle of the "phase rotation". In case of the circuit of FIG. 16(a),after the calculation of angle of the "phase rotation", the compensationof the phase is effected at the time of performing a BPSK delaydetection. Further, FIG. 16(b) illustrates a modification of thereceiving circuit of the tenth embodiment of FIG. 16(a). Thismodification of the receiving circuit controls the oscillation frequencyof a first local oscillator (namely, suppresses increase of theoscillation of the first local oscillator) to make a compensation forthe frequency offset. Further, both of the receiving circuits of FIGS.16(a) and 16(b) can perform the compensation of the frequency offsetover a wider range thereof.

Incidentally, in case of the modification illustrated in FIG. 16(b), thefirst local oscillator 2115 is controlled. However, if a second localoscillator 2119 is controlled instead of the first local oscillator2115, similar effects can be obtained.

(Embodiment 11)

Hereinafter, the eleventh embodiment of the present invention will bedescribed by referring to FIGS. 17(a) and 17(b).

FIG. 17(a) is a schematic block diagram for illustrating theconfiguration of a transmitting circuit of a spread spectrumcommunication system of the DS type according to a further M-PSKmodulation method of the present invention (namely, the eleventhembodiment of the present invention). Further, FIG. 17(b) is a schematicblock diagram for illustrating the configuration of a receiving circuitof the spread spectrum communication system of the DS type of FIG.17(a).

The configuration and operation of the transmitting circuit of FIG.17(a) are similar to those of the transmitting circuit of theconventional system of FIG. 4(a). Thus the descriptions of theconfiguration and operation of the transmitting circuit of FIG. 17(a)are omitted herein for the simplicity of description.

Further, the configuration of the receiving circuit of FIG. 17(b) issimilar to that of the receiving circuit of the tenth embodiment of FIG.16(a). In case of the receiving circuit of FIG. 17(b), when decodingdata, a BPSK detection is first performed on a received signal by usingspreading code signals corresponding to the components I and Q of asignal transmitted by the transmitting circuit. Then, a QPSK decoding isperformed on a result of the BPSK detection. Further, an angle of the"phase rotation" is detected by a phase detecting circuit 2114 by usingan output of a correlator 2120. Then, a phase compensation is effectedby a BPSK detection circuit 2125 as a frequency offset compensation.

Additionally, if the frequency offset compensation is effected bycontrolling the oscillation frequency of the second local oscillators2115 and 2119 according to output signals of by phase detecting circuit2124 (however, such an arrangement is not shown in FIG. 17(b)) as incase of the tenth embodiment of the present invention, similar effectscan be obtained.

(Embodiment 12)

Hereinafter, the twelfth embodiment of the present invention will bedescribed by referring to FIGS. 18(a) and 18(b).

FIGS. 18(a) and 18(b) are diagrams for illustrating a method for afrequency offset compensation. Namely, FIG. 18(a) is a diagram forillustrating the "phase rotation" of an output of a correlatorcorresponding to the spreading code (i) which corresponds to thecomponent I. FIG. 18(b) is a schematic block diagram for illustratingthe configuration of a receiving circuit (namely, a decoding portion) ofa spread spectrum communication system of the DS type for performing aphase compensation.

In FIG. 18(a), X-axis (namely, the horizontal axis) and Y-axis (namely,the vertical axis) are the output V1 of the correlator 2120a and that V3of the correlator 2120b. Further, a point Ri represents the output ofthe correlator corresponding to the spreading code (i). Moreover, anangle θ is an absolute phase angle (namely, angle of phase, which formedby V1-axis (namely, X-axis) and a segment drawing from the origin toRi). The phase detecting devices 2124 of the tenth and eleventhembodiments detect this angle and calculate an amount of the "phaserotation" (namely, the angular displacement). However, if this angle is(π/2), the output V1 becomes equal to "0". Thus it is necessary foreffecting the decoding to use output signals of the other correlatorsaccording to circumstances and on predetermined conditions. To solvethis problem, in case the twelfth embodiment of the present invention,output signals of the correlators of each pair are added together andsignals obtained as the result of the additions are used for decoding.

FIG. 18(b) is a schematic block diagram of the configuration of thereceiving circuit including the correlators 2120a to 2120d and so forth.Further, the twelfth embodiment is different from the ninth embodimentin that adders 2126i and 2126q are provided prior to BPSK delaydetection portions 2122i and 2122q, respectively. Thereby, in case ofthe twelfth embodiment, it becomes unnecessary for decoding to useoutput signals of the correlators according to circumstances and onpredetermined conditions. Moreover, the configuration of the receivingcircuit can be further simplified.

Turning now to FIGS. 19(a) and 19(b), there are shown examples ofcharacteristics of the system in case of the eleventh embodiment of thepresent invention. Further, FIG. 19(a) illustrates the characteristicsof the system in case that the frequency offset occurring in the DQPSKcircuit is not corrected or compensated. In contrast, FIG. 19(b)illustrates the characteristics of the system in case that the frequencyoffset occurring in the DQPSK circuit is corrected or compensated. Ascan be seen from these figures, in accordance with the presentinvention, C/N-BER characteristics are improved in comparison with theconventional system. Further, if the frequency offset compensation isnot effected, the system of the present invention can improve thecharacteristics by nearly one order of each of the values thereof to thefrequency offset of ±15% or so (of what is called the symbol rate).Moreover, as can be seen from FIG. 19(b), when the frequency offsetcompensation is effected, each of the curves representing theBER-characteristics of the embodiment of the present invention becomesalmost flat within the range of ±15% of the symbol rate. This proves theeffectiveness of the present invention. Furthermore, the configurationof the receiving circuit can be simplified.

Incidentally, in each of the transmitting and receiving circuits of theninth to twelfth embodiments of the present invention, two localoscillators (namely, the first and second local oscillators) are used.However, apparently, if the number of local oscillators is changedaccording to the requirement of the communication system, the sameeffect as that of the ninth to twelfth embodiments can be obtained.

(Embodiment 13)

Hereinafter, the thirteenth embodiment will be described by referring toFIGS. 20(a), 20(b), 21 and 22. Incidentally, in case of this embodiment,the QPSK method is employed as an example of the M-QPSK modulationmethod.

FIG. 20(a) is a schematic block diagram for illustrating theconfiguration of a transmitting unit of a DS/DPSK system for performinga spreading code generating method according to the present invention(namely, the thirteenth embodiment of the present invention). FIG. 20(b)is a schematic block diagram for illustrating the configuration of areceiving unit of the DS/DPSK system of FIG. 20(a). For simplicity ofdrawing, a transmission amplifier and a reception front-end portion arenot shown in these figures.

Further, as is shown in FIG. 21, in case of the embodiment of FIGS.20(a) and 20(b), the spreading rate is N (incidentally, N=2^(n) (n is aninteger larger than 2)). Moreover, an orthogonal code used as aspreading code is Hadamard function code having a code length of N_(w)(incidentally, N_(w) =2^(n-2)). Furthermore, PN codes used as spreadingcodes are PNi and PNq, each of which is composed of a code pattern of aperiod of 2^(n-1) and has a code length of N=2^(n). In FIG. 21,reference characters H0 to HN_(w) denote code pattern numbers of the HFcode; and PN₁ to PN_(k) quadrature spreading code sets, each of whichconsists of different code patterns corresponding to the in-phase andquadrature components I and Q, respectively. Additionally, FIG. 22illustrates the code synchronization relation between the PN code andthe HF code of the thirteenth embodiment of the present invention.

First, in FIG. 20(a) showing a transmitting unit 3010, reference numeral3100 designates a spreading code generating portion; 3110 a QPSKencoding circuit for performing a QPSK encoding on input data; 3141 and3142 PN code generators each for generating PN codes, based on clocks;3150 an orthogonal code generator for generating an orthogonal code,based on clocks; 3131 a multiplier for multiplying an orthogonal codeoutputted from the orthogonal code generator 3150 with a PN codeoutputted from the PN code generator 3141; 3132 a multiplier formultiplying the orthogonal code outputted from the orthogonal codegenerator 3150 with a PN code outputted from the PN code generator 3142;3121 a multiplier for multiplying a QPSK code outputted from the QPSKencoding circuit 3110 with an output of the multiplier 3131; 3122 amultiplier for multiplying the QPSK code outputted from the QPSKencoding circuit 3110 with an output of the multiplier 3132; 3160 aquadrature modulation circuit for performing quadrature modulations oneach of output signals of the multipliers 3121 and 3122 according to alocal signal generated from the local signal source 3170; and 3180 aradio antenna for transmitting an output signal of the quadraturemodulation circuit 3160.

In the circuit having the configuration described hereinabove, first, aQPSK encoding is performed on data inputted to the transmitting block orunit 3010 by the QPSK encoding circuit 3110. Then, signal sequencescorresponding to the components or channels Ich and Qch are sent to themultipliers 3121 and 3122, respectively. On the other hand, in thespreading code generating portion 3100, the orthogonal code generator3150 generates HF codes at one-fourth the spreading rate. Moreover, thePN code generators 3141 and 3142 generate different code patterns PNiand PNq at the spreading rate, respectively. Then, the code PNi ismultiplied by the HF code in the multiplier 3131 and further the codePNq is multiplied by the HF code in the multiplier 3132 in the manner asillustrated in FIG. 22. Thus, spreading codes are generated.Subsequently, the data corresponding to the channels Ich and Qch, whichare sent to the multipliers 3121 and 1322, are spread by using thespreading codes generated in the spreading code generating portion 3100.Then, a quadrature modulation is performed on the spread data in thequadrature modulation portion 3160. Thereafter, resultant signals aretransmitted from the radio antenna 3180.

In FIG. 20(b) showing the configuration of the receiving unit 3020,reference numeral 3200 designates a radio antenna for receiving a signaltransmitted from the radio antenna 3180 of FIG. 20(a); 3201 a BPF(namely, a band-pass filter); 3210 a quadrature detection circuit forperforming a quadrature detection on an output signal of the BPF 3201 onthe basis of a local signal generated by a local signal generator 3220;3231 and 3232 LPFs (namely, low-pass filters); 3241 and 3242 A/Dconverters converting analog outputs of the LPFs 3231 and 3232 intodigital signals, respectively; 3251 to 3254 digital correlators forobtaining the correlations of outputs of the A/D converters 3241 and3242 by utilizing what is called a sliding correlation (to be describedlater); 3260 a data decoding circuit for decoding data according tooutputs of the LPFs 3231 and 3232; 3340 a clock recovery circuit forrecovering clocks of the transmitting unit 3010 from outputs of the LPFs3231 and 3232; 3300 a de-spreading code generating circuit consisting ofcomposing elements designated by the following reference numerals.

Namely, reference numerals 3281 and 3282 denote PN code generators forgenerating PN codes according to outputs of the clock recovery circuit3340 and a synchronization acquisition circuit (to be described later);3290 an orthogonal code generator for generating an orthogonal codeaccording to outputs of the clock recovery circuit 3340 and thesynchronization acquisition circuit (to be described later); and 3271and 3272 modulo-2 multipliers multiplying an orthogonal code outputtedby the orthogonal code generator 3290 with the PN codes (namely, PNi andPNq) of the PN code generators 3281 and 3282, respectively.

Meanwhile, reference numerals 3311 and 3312 are squaring devicescalculating the squares of output signals of the correlators 3251 and3253, respectively; 3320 an adder for adding up outputs of the squaringdevices 3311 and 3312; and 3330 the synchronization acquisition circuitfor performing a synchronization acquisition judgment from an output ofthe adder 3320.

In the receiving unit 3020 of FIG. 20(b) having the configurationdescribed hereinabove, a DS/QPSK modulation signal generated andtransmitted by the transmitting unit 3010 of FIG. 20(a) is received bythe radio antenna 3200. Then, a quadrature detection is performed byusing a local signal sent from the local signal source 3220 on thereceived signal inputted to the quadrature detection circuit 3210through the BPF 3201. Subsequently, signals corresponding to thechannels Ich and Qch are outputted from the quadrature detection circuit3210 in parallel with each other. Further, the signals corresponding tothe channels Ich and Qch are inputted through the LPFs 3231 and 3232 tothe A/D converters 3241 and 3242, whereupon A/D conversions areperformed on signals inputted thereto. Thereafter, outputs of the A/Dconverters 3241 and 3242 are inputted to the digital correlators 3251 to3254, as illustrated in FIG. 20(b).

In case of this embodiment, each of the digital correlators 3251 to 3254calculates what is called a sliding correlation. Further, in order tocope with the "phase rotation" occurring due to the frequency offset,these four correlators are used. Moreover, a synchronization acquisitionjudgment is performed by the synchronization acquisition judgmentcircuit 3330 by using the squares of outputs of the correlators 3251 and3253. Namely, the code patterns are circulated and are synchronized witheach other in case where the cross correlation therebetween becomesminimum.

Further, the de-spreading code generating circuit 3300 is similar to thespreading code generating portion 3100 of the transmitting unit 3010,except that the synchronization acquisition circuit 3330 controls thetiming of each de-spreading code generation.

Incidentally, in case of this embodiment, the code length of the PN codeis the quadruple of the code length of the HF code. Further, the clockrates of input clocks to the orthogonal code generator 3150 and 3290 arereduced by a factor of four by using the clock number converters 3150Aand 3290A, respectively. However, as long as N≧N_(w), any other ratiofor changing the clock rate of an input clock may be employed.

As the result of constructing the transmitting unit 3010 and thereceiving unit 3020 as above described, the spreading code can begenerated by utilizing the characteristics of the PN codes and the HFcode. Moreover, the degree of multiplexing can be increased.

(Embodiment 14)

Hereinafter, the fourteenth embodiment of the present invention, whichemploys the DS/QPSK method, will be described by referring to FIGS. 23to 25.

FIG. 23 is a diagram for illustrating patterns of codes employed in theDS/QPSK system for performing a spreading code generating methodaccording to the present invention (namely, the fourteenth embodiment ofthe present invention) in case where a 7-stage M-code having a codelength of 127 is employed. Namely, this figure shows the code patternsobtained by using 9 symbols or elements C₁ to C₉ of the 7-stage M-code.Further, the state of the code pattern of this figure is assumed to be astate in which there is no phase offset (namely, all of initial valuesof registers of the code generator).

FIG. 25 is a diagram for illustrating maximum values of the crosscorrelations between the code patterns of FIG. 23. As is seen from FIG.25, a maximum value of the cross correlation of a code pattern varieswith the code pattern (see the maximum values of, for instance, thecross correlation between elements C₁ and C₅ and the cross correlationbetween elements C₁ and C₇ of FIG. 25) even if the code patterns areobtained from elements of the same M-code. Thus, the code patterns(namely, the combinations of elements, for example, (C₁ -C₅), (C₂ -C₆),(C₃ -C₇), (C₄ -C₈)) having large maximum values of the correlationsthereof are used as quadrature spreading code sets corresponding to thein-phase channel Ich and the quadrature channel Qch. In case of each ofthe code patterns or sets, a phase offset is caused in one of theelements thereof such that the cross correlation of each of the codesets or patterns has a minimum value. FIG. 24 is a diagram forillustrating practical examples of quadrature spreading code sets orpatterns generated in the fourteenth embodiment of the presentinvention. For example, in case of the quadrature spreading code set (C₁-C₅), phase offsets are caused in seven chip (codes) of the element C₅and as a consequence the cross correlation between these two elementsbecomes equal to 1.

As described above, in case of this embodiment, code patterns havinglarge cross correlations thereof are used as quadrature spreading codes.Further, an phase offset is caused in the code patterns. Thereby, evenif the in-phase channel or component Ich is not completely separatedfrom the quadrature component Qch in the receiving unit or station, theinterference between these components can be made to be small. Moreover,the cross correlation therebetween can be made to be relatively small.Furthermore, even when one of the channels is interfered, the otherchannel can be detected. Consequently, in comparison with theconventional system, the possibility of an occurrence of malfunction canbe reduced.

While the preferred embodiments of the present invention have beendescribed above, it is to be understood that the present invention isnot limited thereto and that other modifications will be apparent tothose skilled in the art without departing from the spirit of theinvention. The scope of the present invention, therefore, is to bedetermined solely by the appended claims.

What is claimed is:
 1. An M-phase shift keying modulation method (M isan integer greater than 2) for use in a direct sequence type of spreadspectrum communication system having a transmitting unit and a receivingunit, the M-phase shift keying modulation method comprising the stepsof:producing an in-phase symbol signal denoting a data sequence of anin-phase symbol and a quadrature symbol signal denoting a data sequenceof a quadrature symbol from an input signal in the transmitting unit;spreading a spectrum of the in-phase symbol signal to obtain an in-phasespread spectrum signal by multiplying the in-phase symbol signal by aspreading code signal of the in-phase symbol in the transmitting unit,the spreading code signal of the in-phase symbol having a firstspreading code; spreading a spectrum of the quadrature symbol signal toobtain a quadrature spread spectrum signal by multiplying the quadraturesymbol signal by a spreading code signal of the quadrature symbol in thetransmitting unit, the spreading code signal of the quadrature symbolhaving a second spreading code different from the first spreading code;performing a quadrature modulation for the in-phase spread spectrumsignal and the quadrature spread spectrum signal in the transmittingunit to obtain a synthesized signal; mixing the synthesized signal witha first local oscillating signal to convert a frequency band of thesynthesized signal into a carrier frequency band; transmitting thesynthesized signal of the carrier frequency band from the transmittingunit to the receiving unit through a transmission path; mixing thesynthesized signal with a second local oscillating signal in thereceiving unit to convert the carrier frequency band of the synthesizedsignal into an intermediate frequency band; performing a quadraturedetection for the synthesized signal of the intermediate frequency bandin the receiving unit to demodulate the synthesized signal to a firstdata signal and a second data signal; correlating the first data signalwith the spreading code signal of the in-phase symbol in the receivingunit to obtain a first in-phase correlated signal; correlating the firstdata signal with the spreading code signal of the quadrature symbol inthe receiving unit to obtain a first quadrature correlated signal;correlating the second data signal with the spreading code signal of thein-phase symbol in the receiving unit to obtain a second in-phasecorrelated signal; correlating the second data signal with the spreadingcode signal of the quadrature symbol in the receiving unit to obtain asecond quadrature correlated signal; detecting an angle of a phaserotation between the first in-phase correlated signal and the secondin-phase correlated signal in the receiving unit, the angle of the phaserotation indicating a phase shift between the first and second in-phasecorrelated signals; adjusting an oscillating frequency of the secondlocal oscillating signal according to the angle of the phase rotation tocompensate the phase shift between the first and second in-phasecorrelated signals; performing a binary phase shift keying detection forthe first and second in-phase correlated signals in the receiving unitto reproduce the in-phase symbol signal; performing the binary phaseshift keying detection for the first and second quadrature correlatedsignals in the receiving unit to reproduce the quadrature symbol signal;and decoding the in-phase symbol signal and the quadrature symbol signalin the receiving unit to reproduce the input signal.
 2. An M-phase shiftkeying modulation method according to claim 1 in which the step ofperforming a binary phase shift keying detection for the first andsecond in-phase correlated signals comprises the steps of:adding thefirst in-phase correlated signal and the second in-phase correlatedsignal to produce a first summed signal; and performing a binary phaseshift keying detection for the first summed signal to reproduce thein-phase symbol signal, and the step of performing the binary phaseshift keying detection for the first and second quadrature correlatedsignals comprises the steps of:adding the first quadrature correlatedsignal and the second quadrature correlated signal to produce a secondsummed signal; and performing the binary phase shift keying detectionfor the second summed signal to reproduce the quadrature symbol signal.3. An M-phase shift keying modulation method (M is an integer greaterthan 2) for use in a direct sequence type of spread spectrumcommunication system having a transmitting unit and a receiving unit,the M-phase shift keying modulation method comprising the stepsof:producing an in-phase symbol signal denoting a data sequence of anin-phase symbol and a quadrature symbol signal denoting a data sequenceof a quadrature symbol from an input signal in the transmitting unit;performing a differential encoding for the in-phase symbol signalaccording to a differential binary phase shift keying modulation in thetransmitting unit to obtain an in-phase differential code signaldenoting a differential code sequence of the in-phase symbol; performingthe differential encoding for the quadrature symbol signal according tothe differential binary phase shift keying modulation in thetransmitting unit to obtain a quadrature differential code signaldenoting a differential code sequence of the quadrature symbol;spreading a spectrum of the in-phase differential code signal to obtainan in-phase spread spectrum signal by multiplying the in-phasedifferential code signal by a spreading code signal of the in-phasesymbol in the transmitting unit, the spreading code signal of thein-phase symbol having a first spreading code; spreading a spectrum ofthe quadrature differential code signal to obtain a quadrature spreadspectrum signal by multiplying the quadrature differential code signalby a spreading code signal of the quadrature symbol in the transmittingunit, the spreading code signal of the quadrature symbol having a secondspreading code different from the first spreading code; performing aquadrature modulation for the in-phase spread spectrum signal and thequadrature spread spectrum signal in the transmitting unit to obtain asynthesized signal; transmitting the synthesized signal from thetransmitting unit to the receiving unit; performing a quadraturedetection for the synthesized signal in the receiving unit to demodulatethe synthesized signal to a first data signal and a second data signal;correlating the first data signal with the spreading code signal of thein-phase symbol in the receiving unit to obtain a first in-phasecorrelated signal; correlating the first data signal with the spreadingcode signal of the quadrature symbol in the receiving unit to obtain afirst quadrature correlated signal; correlating the second data signalwith the spreading code signal of the in-phase symbol in the receivingunit to obtain a second in-phase correlated signal; correlating thesecond data signal with the spreading code signal of the quadraturesymbol in the receiving unit to obtain a second quadrature correlatedsignal; performing a binary phase shift keying detection for the firstand second in-phase correlated signals in the receiving unit toreproduce the in-phase differential code signal; performing a binaryphase shift keying detection for the first and second quadraturecorrelated signals in the receiving unit to reproduce the quadraturedifferential code signal; decoding the in-phase differential code signaland the quadrature differential code signal in the receiving unit toreproduce the in-phase symbol signal and the quadrature symbol signal;and reproducing the input signal from the in-phase symbol signal and thequadrature symbol signal.
 4. An M-phase shift keying modulation methodaccording to claim 3 in which the step of transmitting the synthesizedsignal comprises the steps of:mixing the synthesized signal with a firstlocal oscillating signal to convert a frequency band of the synthesizedsignal into a carrier frequency band; transmitting the synthesizedsignal of the carrier frequency band from the transmitting unit to atransmission path; receiving the synthesized signal of the carrierfrequency band transmitting through the transmission path in thereceiving unit; and mixing the synthesized signal with a second localoscillating signal to convert the carrier frequency band of thesynthesized signal into an intermediate frequency band, and the M-phaseshift keying modulation method further comprising the steps of:detectingan angle of a phase rotation between the first in-phase correlatedsignal and the second in-phase correlated signal; and adjusting anoscillating frequency of the second local oscillating signal accordingto the angle of the phase rotation to compensate a phase shift betweenthe first and second in-phase correlated signals.
 5. An M-phase shiftkeying modulation method according to claim 3, further comprising thesteps of:detecting an angle of a phase rotation between the firstin-phase correlated signal and the second in-phase correlated signal;compensating the binary phase shift keying detection for a first phaseshift between the first and second in-phase correlated signals accordingto the angle of the phase rotation; and compensating the binary phaseshift keying detection for a second phase shift between the first andsecond quadrature correlated signals according to the angle of the phaserotation.
 6. An M-phase shift keying modulation method according toclaim 3 in which the step of performing a binary phase shift keyingdetection for the first and second in-phase correlated signals comprisesthe steps of:adding the first in-phase correlated signal and the secondin-phase correlated signal to produce a first summed signal; andperforming a binary phase shift keying detection for the first summedsignal to reproduce the in-phase differential code signal, and the stepof performing the binary phase shift keying detection for the first andsecond quadrature correlated signals comprises the steps of:adding thefirst quadrature correlated signal and the second quadrature correlatedsignal to produce a second summed signal; and performing the binaryphase shift keying detection for the second summed signal to reproducethe quadrature differential code signal.
 7. An M-phase shift keyingmodulation method (M is an integer greater than 2) for use in a directsequence type of spread spectrum communication system having atransmitting unit and a receiving unit, the M-phase shift keyingmodulation method comprising the steps of:producing an in-phase symbolsignal denoting a data sequence of an in-phase symbol and a quadraturesymbol signal denoting a data sequence of a quadrature symbol from aninput signal in the transmitting unit; spreading a spectrum of thein-phase symbol signal to obtain an in-phase spread spectrum signal bymultiplying the in-phase symbol signal by a spreading code signal of thein-phase symbol in the transmitting unit, the spreading code signal ofthe in-phase symbol having a first spreading code; spreading a spectrumof the quadrature symbol signal to obtain a quadrature spread spectrumsignal by multiplying the quadrature symbol signal by a spreading codesignal of the quadrature symbol in the transmitting unit, the spreadingcode signal of the quadrature symbol having a second spreading codedifferent from the first spreading code; performing a quadraturemodulation for the in-phase spread spectrum signal and the quadraturespread spectrum signal in the transmitting unit to obtain a synthesizedsignal; transmitting the synthesized signal from the transmitting unitto the receiving unit; performing a quadrature detection for thesynthesized signal in the receiving unit to demodulate the synthesizedsignal to a first data signal and a second data signal; correlating thefirst data signal with the spreading code signal of the in-phase symbolin the receiving unit to obtain a first in-phase correlated signal;correlating the first data signal with the spreading code signal of thequadrature symbol in the receiving unit to obtain a first quadraturecorrelated signal; correlating the second data signal with the spreadingcode signal of the in-phase symbol in the receiving unit to obtain asecond in-phase correlated signal; correlating the second data signalwith the spreading code signal of the quadrature symbol in the receivingunit to obtain a second quadrature correlated signal; detecting an angleof a phase rotation between the first in-phase correlated signal and thesecond in-phase correlated signal, the angle of the phase rotationindicating a first phase shift between the first and second in-phasecorrelated signals and a second phase shift between the first and secondquadrature correlated signals; performing a binary phase shift keyingdetection for the first and second in-phase correlated signals in thereceiving unit to reproduce the in-phase symbol signal whilecompensating the first phase shift between the first and second in-phasecorrelated signals according to the angle of the phase rotation;performing the binary phase shift keying detection for the first andsecond quadrature correlated signals in the receiving unit to reproducethe quadrature symbol signal while compensating the second phase shiftbetween the first and second quadrature correlated signals according tothe angle of the phase rotation; and decoding the in-phase symbol signaland the quadrature symbol signal in the receiving unit to reproduce theinput signal.
 8. An M-phase shift keying modulation method according toclaim 7 in which the step of performing a binary phase shift keyingdetection for the first and second in-phase correlated signals comprisesthe steps of:adding the first in-phase correlated signal and the secondin-phase correlated signal to produce a first summed signal; andperforming a binary phase shift keying detection for the first summedsignal to reproduce the in-phase symbol signal while compensating thefirst phase shift between the first and second in-phase correlatedsignals according to the angle of the phase rotation, and the step ofperforming a binary phase shift keying detection for the first andsecond quadrature correlated signals comprises the steps of:adding thefirst quadrature correlated signal and the second quadrature correlatedsignal to produce a second summed signal; and performing the binaryphase shift keying detection for the second summed signal to reproducethe quadrature symbol signal while compensating the second phase shiftbetween the first and second quadrature correlated signals according tothe angle of the phase rotation.